The mobile TV reception front end must be designed to maintain high sensitivity when operating far from the transmitter and also withstand overload in the presence of strong signals. It is intended for integration into in-vehicle entertainment (ICE) systems as well as portable devices like smartphones, PDAs, and laptops. The system should perform reliably even when the receiver is at a distance from the transmitter and under varying conditions, which differs from traditional broadcast TV. A cost-effective solution involves combining a high-gain, low-noise amplifier (LNA) with a PIN diode bypass switch, offering both overload protection and high sensitivity.
A table (Table 1) illustrates the gain and IP3 performance of the receiver front end shown in Figure 1. The most practical implementation of a mobile TV receiver involves reducing the gain under strong signal conditions. This approach simplifies the linearity requirements of the mixer stage, allowing the use of cost-effective RF ICs. In a cascade analysis with a variable gain receiver front end, the improvement in input third-order intercept point (IIP3) depends on the gain variation (as shown in Figure 1 and Table 1). Adjustable gain receivers are more effective at handling strong signals compared to fixed gain designs.
An Automatic Gain Control (AGC) circuit can adjust the LNA gain, and since AGC is typically placed before the channel filter, it can respond to overloads from adjacent channels. One method to reduce RF gain is by shunting part of the signal to ground before the LNA, as shown in Figure 2(a). While this method uses fewer RF switches, it may cause impedance mismatch when the switch is off. An alternative is to connect a damper element to the high-impedance or "hot" end of the LNA's parallel resonant network, though this reduces RF selectivity.
When the received signal overloads the stages after the LNA, such as the mixer or IF amplifier, the LNA can be bypassed using RF switches. In bypass mode, the input signal is sent directly to the downconverter IC, as shown in Figure 2(b). As long as the components match the characteristic impedance (75Ω for mobile TV), mismatch issues are minimized. However, adding switches increases circuit complexity.
Another technique to reduce RF gain is by lowering the quiescent current supplied to the LNA, as seen in Figure 2(c). Devices like dual-gate MOSFETs use additional terminations to control bias current. This method avoids switching elements but sacrifices some linearity due to lower collector/drain current.
To meet the requirements of dual-mode (analog/digital) mobile TV receivers across the 47–870 MHz spectrum, several MMIC options were evaluated, but their linearity was insufficient. A broadband high-linearity MMIC LNA (MGA-68563) combined with an external PIN diode switch was chosen for the design.
This single-stage GaAs PHEMT LNA device has a gate width of 800 microns (Figure 3). Its gate is connected to an internal current mirror to compensate for process variations and minimize threshold voltage effects. The LNA uses lossy negative feedback to ensure stability and smooth amplitude response within a 3dB window (±1.5dB) across 100MHz–1GHz.
The MMIC does not require output impedance matching due to its internal feedback and less than 10dB output return loss. However, input matching across such a wide frequency range (47–870MHz) proved challenging. To optimize input return loss, the drain current (Ids) was set to 30mA, higher than the nominal 10mA, to account for potential issues with the PIN diode switching circuitry. Pin 4 controls the current through an external resistor R1, adjusting Ids while keeping the supply voltage Vd at 3V. Increasing Ids by three times improves linearity.
In designing the LNA/switch circuit, the original configuration used four PIN diodes for a DPDT switch (Figure 5a). However, this design was complex, so a simpler DPST switch (Figure 5b) was developed. In bypass mode, the LNA power supply (Vdd) is turned off to utilize the inherent isolation of the unbiased FET. This approach slightly degrades return loss due to finite gate and drain impedances.
Inductors L1 and L2 act as ferrite beads, providing high impedance over the entire MMIC and diode bias network (Figure 5b). Without L1, part of the input signal could bypass to ground through parasitic capacitance. Capacitors C3, C4, and C5 decouple RF signals from DC sources, with reactance values of about 5Ω at the lowest frequency. C1 and C2 block DC at the MMIC input and output, with C2 chosen to provide a high-pass response that compensates for high-frequency gain roll-off.
Resistors R1 and R2 control the MMIC current, setting it to 30mA at Vdd = 3V. At VSW = 3V, resistors R3, R4, and R5 limit the forward bias of the PIN diode to ~2.5mA. Using one PIN diode instead of two offers no benefit, as the footprint and cost are similar.
A prototype was built on a PCB using Rogers RO4350B laminate. At 10 GHz, the dielectric constant is 3.48. The PIN diode and biasing components were soldered to existing pads. Axial glass diodes (1N5719) were used initially, later replaced with SOT-packaged PIN diode pairs (HSMP-3893/E).
The median gain of the LNA is 19.8 dB ± 1.3 dB in the frequency range of interest (Figure 6a). The high-pass response of C2 moderately attenuates signals below 200MHz, ensuring a flat frequency response. High-frequency gain roll-off aligns with MMIC characteristics, possibly due to parasitic capacitance in the unbiased PIN diode.
In bypass mode, the circuit exhibits 3.8–4.5 dB attenuation across the spectrum (Figure 6a). Loss is mainly due to the parasitic series inductance of the PIN diode. PCB dissipation, FET termination impedance, and R4’s shunt capacitance also contribute, but the loss remains within the customer’s -5 dB specification. Efforts are ongoing to further reduce bypass loss.
Input and output return loss performance is consistently good (less than 17 dB) in bypass mode. The proximity of the unbiased FET’s gate and drain to the open-loop circuit affects return loss. When the LNA is active, output return loss at the lowest frequency reaches 7 dB, caused by the small value of C2, which was chosen to improve frequency response.
Figure 7a compares the LNA noise figure with and without the ferrite bead inductor L1. Without L1, the target noise level (≤1.3dB) cannot be met. Parasitic capacitance of R3 adds 0.3–0.6dB to the noise. With L1, the noise changes slightly (from 0.2dB to 0.5dB), but this is minor. These changes may be due to increased turbulence in the ferrite bead at higher frequencies, especially near its self-resonant frequency (~100MHz).
In the mobile TV frequency band, the LNA’s output third-order intercept point (OIP3) was measured at uniformly distributed points using a two-tone input of -20 dBm. IIP3 was calculated by subtracting the measured gain from OIP3 data. The OIP3 is ≥30.3 dBm, with a maximum gain variation of 0.8 dB (Figure 7b). There is a 10dB improvement over the nominal 20dBm value from the datasheet, attributed to the higher Ids used in the design.
The LNA/switch design meets its specifications and shows significant potential for further improvement, such as enhanced noise performance by using higher SRF ferrite bead inductors.
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